Envelope modulator

ABSTRACT

In an embodiment an envelope modulator comprises a gain compensated lag-lead network or a gain compensated lead-lag network.

FIELD

Embodiments described herein relate generally to envelope modulators,and in particular to split frequency envelope modulators.

BACKGROUND

The technique of envelope modulation is used to increase the efficiencyof radio frequency power amplifiers. The supply voltage to the poweramplifier is modulated according to the envelope signal of the radiofrequency signal. In a split frequency envelope modulator, the envelopesignal is split into two parts: a low frequency or DC component and ahigh frequency or AC component. These two components are amplifiedseparately. The split frequency envelope modulator architectureincreases the efficiency of envelope modulation; however, spiltfrequency envelope modulators suffer from a phase transient at thecross-over frequency between the low frequency component and the highfrequency component.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments will be described as non-limiting examples with reference tothe accompanying drawings in which:

FIG. 1a shows an envelope modulator;

FIG. 1b shows the frequency response of the envelope modulator shown inFIG. 1 a;

FIGS. 2a and 2b show lag-lead networks;

FIG. 2c shows the frequency response of the lag-lead networks shown inFIGS. 2a and 2 b;

FIG. 3 shows a gain compensated lag-lead network;

FIG. 4a shows an envelope modulator according to an embodiment;

FIG. 4b shows the frequency response of the modulator shown in FIG. 4 a;

FIG. 5 shows an envelope modulator according to an embodiment;

FIG. 6 shows a synthetic impedance circuit;

FIGS. 7a and 7b show lead-lag networks;

FIG. 7c shows the frequency response of the lead-lag networks shown inFIGS. 7a and 7 b;

FIG. 8 shows a gain compensated lead-lag network;

FIG. 9a shows an envelope modulator according to an embodiment;

FIG. 9b shows the frequency response of the modulator shown in FIG. 9 a;

FIG. 10 shows an envelope modulator according to an embodiment; and

FIG. 11 shows a power amplifier circuit according to an embodiment.

DETAILED DESCRIPTION

In one embodiment, an envelope modulator comprises a gain compensatedlag-lead network or a gain compensated lead-lag network.

In an embodiment, the envelope modulator comprises a low frequency pathconfigured to amplify components of an input signal having frequenciesbelow a first threshold and a high frequency path configured to amplifycomponents of the input signal having frequencies above a secondthreshold.

In an embodiment, the first threshold is greater than the secondthreshold.

In an embodiment, the first threshold is at least five times greaterthan the second threshold.

In an embodiment, the modulator comprises a lag-lead network and thehigh frequency path comprises a first impedance and a first amplifierconfigured to provide gain compensation for the first impedance.

In an embodiment, the first impedance is a synthetic impedance.

In an embodiment, the modulator comprises a lead-lag network and the lowfrequency path comprises a second impedance and a second amplifierconfigured to provide gain compensation for the second impedance.

In an embodiment, the second impedance is a synthetic impedance.

In an embodiment, the second amplifier is a switched mode power supply.

In one embodiment a power amplifier circuit comprises a power amplifierconfigured to amplify an input radiofrequency signal and an envelopemodulator as described above and configured to modulate a power supplyof the power amplifier according to an envelope of the inputradiofrequency signal.

FIG. 1a shows an envelope modulator 10. The modulator 10 is a splitfrequency modulator comprises a low frequency path 12 and a highfrequency path 14 connected in parallel. An input signal is coupled toboth the low frequency path 12 and the high frequency path 14 at aninput terminal 13.

The low frequency path 12 comprises a switched mode power supply SMPS. Afirst inductor L2 is connected in the low frequency path 12 between theinput terminal 13 and the switched mode power supply SMPS. A resistor R4is connected between the input of the switched mode power supply SMPSand ground. The first inductor L2 and the resistor R4 together form aninput low pass filter. A second inductor L3 is connected between theoutput of the switched mode power supply SMPS and a load R5.

The high frequency path 14 comprises an amplifier G1. A first capacitorC2 is connected in the high frequency path 14 between the input terminal13 and the amplifier G1. A resistor R3 is connected between the input ofthe amplifier G1 and ground. The first capacitor C2 and the resistor R3form an input high pass filter. A second capacitor C3 is connectedbetween the output of the amplifier G1 and the load R5.

In use, the modulator 10 modulates the supply voltage of a poweramplifier according to an envelope signal. The switched mode powersupply SMPS in the low frequency path 12 amplifies DC and low frequencycomponents of the envelope signal. The amplifier G1 in the highfrequency path amplifies high frequency components or AC components ofthe envelope signal.

When the modulator is used in an envelope modulated radiofrequencyamplifier the load R5 represents the power amplifier. It is assumed thatthe equivalent impedance of the power amplifier is known, and hence canbe represented by a resistor.

FIG. 1b shows the frequency response of the envelope modulator shown inFIG. 1a . As shown in FIG. 1b , the gain response is flat with respectto frequency. This is based on the assumption that both the inputfilters and output filters of both the low frequency path and the highfrequency path have the same cut-off frequency. However, as shown inFIG. 1b , there is a large phase transient at the cross-over frequency.At frequencies below the cross-over frequency the output signal isdominated by the response of the low frequency path. At frequenciesabove the cross-over frequency the output signal is dominated by theresponse of the high frequency path.

The phase transient shown in FIG. 1b where the phase flips from −180degrees to +180 degrees is due to the low frequency path and the highfrequency path working against each other in the cross-over region.

FIG. 2a shows a lag-lead network 20. The lag-lead network 20 is aresistor-capacitor-resistor lag-lead network. The lag-lead network 20comprises a first resistor R1, a second resistor R2 and a capacitor C1.The first resistor R1 is connected between an input terminal 23 and anoutput terminal 25. The second resistor R2 and the capacitor C1 areconnected in series between the output terminal 25 and ground.

FIG. 2b shows a lag-lead network 21. The lag-lead network 21 is theresistor-inductor-resistor equivalent of the lag-lead network 20 shownin FIG. 2a . The resistor-inductor-resistor lag-lead network 21comprises a first resistor R1, an inductor L1 and a second resistor R2.The first resistor R1 and the inductor L1 are connected in parallelbetween an input terminal 26 and an output terminal 27. The secondresistor R2 is connected between the output terminal 27 and ground. Thelag-lead network 21 may be considered to have a low frequency path 22and a high frequency path 24. The low frequency path 22 is formed by theinductor L1 which has low impedance for low frequency signals but highimpedance for high frequency signals. The high frequency path 24 isformed by the resistor R1 which for high frequency signals has lowerimpedance than the inductor L1.

The lag-lead networks shown in FIGS. 2a and 2b are low pass lag-leadnetworks.

FIG. 2c shows the frequency response of the resistor-capacitor-resistorlag-lead network 20 shown in FIG. 2a and the equivalentresistor-inductor-resistor lag-lead network 21 shown in FIG. 2b . Bothof the lag-lead networks have the frequency response shown in FIG. 2 c.

As shown in FIG. 2c , the lead lag networks 20 & 21 attenuate highfrequency signals having a frequency greater than F2 more than lowfrequency signals having a frequency less than F1. This can beunderstood by considering the resistor-capacitor-resistor lag-leadnetwork 20 shown in FIG. 2a . For low frequencies the capacitor C1 hashigh impedance therefore the path from the input terminal 23 to theoutput terminal 25 dominates over the path from the input terminal 23 toground through the capacitor C1. For high frequency signals a portion ofthe input signal will pass through the path to ground via the secondresistor R2 and the capacitor C1. Therefore the gain for the lag-leadnetwork 20 is lower for high frequency signals than for low frequencysignals. The attenuation of high frequency signals can also beunderstood by considering the resistor-inductor-resistor lag-leadnetwork 21 shown in FIG. 2b . For low frequencies the impedance of theinductor L1 is low and therefore low frequency signals are notattenuated. For high frequency signals the high frequency path includingthe resistor R1 dominates, however the resistor R1 does attenuate thehigh frequency signals. As shown in FIG. 2c , the phase response isrelatively flat.

FIG. 3 shows a gain compensated lag-lead network 30. The gaincompensated lag-lead network 30 is based on theresistor-inductor-resistor lag-lead network 21 shown in FIG. 2b . Thegain compensated lag-lead network 30 comprises a low frequency path 32and a high frequency path 34 which are connected in parallel between aninput terminal 33 and an output terminal 35. The low frequency path 32comprises an inductor L1. The high frequency path 34 includes anamplifier G1 connected in series between the input terminal 33 and afirst resistor R1. A second resistor R2 is connected between the outputterminal 27 and ground.

By including the amplifier G1 before the first resistor R1 the highfrequency gain can be matched to the low frequency gain. The gain of G1depends on the ratio of R1 to R2 which are set according to thefollowing formula if equal gain is required for both high and lowfrequencies:

$\frac{G\; 1 \times R\; 2}{{R\; 1} + {R\; 2}} = 1$

The gain compensated lag-lead network 30 has a flat gain and phaseresponse.

FIG. 4a shows an envelope modulator 40 according to an embodiment. Theenvelope modulator 40 comprises a low frequency path 42 and a highfrequency path 44. The low frequency path 42 and the high frequency path44 are connected in parallel between an input terminal 43 and a load R2.

The low frequency path 42 comprises a resistor R4, a switched mode powersupply SMPS and an inductor L1 connected in series. A capacitor C4 isconnected between the input of the switched mode power supply SMPS andground. The capacitor C4 and the resistor R4 form a low pass filter.

The high frequency path 44 comprises an amplifier G1, a resistor R1 anda capacitor C3 connected in series between the input terminal 43 and theload R2.

Comparing FIG. 3 and FIG. 4a , it can be seen that the envelopemodulator 40 shown in FIG. 4a may be considered to be the gaincompensated lag-lead network 30 shown in FIG. 3 with additionalcomponents added. The additional components are the capacitor C3 in thehigh frequency path, and in the low frequency path, the switched modepower supply SMPS and the low pass filter formed from the resistor R4and the capacitor C4.

Adding capacitor C3 to gain compensated lag-lead network shown in FIG. 3to form the envelope modulator 40 shown in FIG. 4a will not greatlychange the frequency response if C3 is large enough. Similarly, addingan SMPS and low pass filter composed of R4 and C4 has little effect iftheir cut-off frequency is above that of L1 (which is equivalent to L3in FIG. 1a ) and the load resistance R2 (equivalent to R5 in FIG. 1a ).In FIG. 4a , C3 will form a high pass filter with R1 and R2. It isassumed here that the cut-off frequency of this filter is lower than thecut-off frequency formed by L1 and R2. For example, the cut-offfrequency formed by C3, R1 and R2 could be five times lower than that ofL1 and R2.

FIG. 4b shows the frequency response of the modulator shown in FIG. 4a .Here it is assumed for simplicity that R1=R2. If R1 is equal to R2, thegain distortion is 7 dB and the phase distortion is 50°. However thisdoes not have to be the case and different values for R1 and R2 may beused.

The addition of R1 in the modulator shown in FIG. 4a improves thefrequency response provided G1 has an appropriate gain. Increasing thevalue of R1 beyond R2 will improve the frequency response further, butthe extra loss associated with this will decrease the efficiency of thesplit frequency envelope modulator (SFEM). By including the resistor R1the current flowing between the two paths is reduced.

In the applications envisaged for the envelope modulators described thecut off frequency is around 15 kHz. Depending on the architecture thecut off frequency could be anywhere between 100 Hz and 10 MHz.

FIG. 5 shows an envelope modulator 50 according to an embodiment. Theenvelope modulator 50 comprises a low frequency path 52 and a highfrequency path 54 which are connected in parallel between an inputterminal 53 and a load R2. The low frequency path 52 is the same as thelow frequency path 42 described above in relation to FIG. 4a . The highfrequency path 54 includes a synthetic impedance in place of theresistor R1 in FIG. 4a . A synthetic impendence is an impedance that issimulated by the feedback around an amplifier. The synthetic impedanceis formed by a resistor R6, an amplifier G2 and the positive andnegative feedback.

As shown in FIG. 5, the high frequency path comprises an amplifier G2,the resistor R6 and a capacitor C3 connected in series. Two feedbackloops are connected to the input of the amplifier G2. A first feedbackloop 57 is connected to the output of the amplifier before the resistorR6. The feedback signal of the first feedback loop is subtracted fromthe input of the amplifier G2. A second feedback loop 58 is connected toa point between the synthetic impedance R6 and the capacitor C3. Thefeedback signal of the second feedback loop is added to the input to theamplifier G2.

R6 has a significantly lower value that R1. The feedback around theamplifier G2 makes the resistance of R6 appear to have a higher value.The use of a synthetic impedance improves the efficiency of the envelopemodulator 50.

FIG. 6 shows an example of a synthetic impedance circuit 60 which may beused in embodiments. The synthetic impedance circuit 60 comprises twooperational amplifiers 61 & 62 and seven resistors R6-R12 which formconnections between the operational amplifiers and an input terminal 63and an output terminal 64.

Those of skill in the art will appreciate that in embodiments,alternative synthetic impedance circuits to that shown in FIG. 6 may beused.

In the embodiments described above the high frequency path, or the ACpath of an envelope modulator includes additional elements.Alternatively, in embodiments the additional elements are included inthe low frequency, or DC path. Such embodiments are described below.

FIG. 7a shows a lead-lag network 70. The lead-lag network 70 comprises afirst resistor R1, a second resistor R2 and an inductor L4. The firstresistor R1 is connected between an input terminal 73 and an outputterminal 75. The second resistor and the inductor are connected inseries between the output terminal 75 and ground.

FIG. 7b shows a lead-lag network 71. The lead-lag network 71 shown inFIG. 7b is the capacitor equivalent of the high pass lag-lead network 70shown in FIG. 7a . The lead-lag network 71 comprises a first resistorR1, a second resistor R2 and a capacitor C5. The first resistor R1 andthe capacitor C5 are connected in parallel between an input terminal 76and an output terminal 77. The second resistor R2 is connected betweenthe output terminal 77 and ground. The lead-lag network 71 may beconsidered to comprise a low frequency path 72 and a high frequency path74. The low frequency path 72 is through the first resistor R1. The highfrequency path 74 is through the capacitor C5. For low frequencysignals, the impedance of the capacitor C5 is high so the path throughthe first resistor R1 dominates. For high frequency signals, theimpedance of the capacitor C5 is low so the path through the capacitorC5 dominates.

FIG. 7c shows the frequency response of both the lead-lag network 70shown in FIG. 7a and the lead-lag network 71 shown in FIG. 7 b.

As shown in FIG. 7c , the lead-lag networks 70 & 71 attenuate highfrequency signals having a frequency greater than F2 less than lowfrequency signals having a frequency less than F1. This can beunderstood by considering the resistor-capacitor-resistor network 71shown in FIG. 7b . For high frequencies the impedance of the capacitorC5 is low and therefore high frequency signals are not attenuated. Forlow frequency signals the low frequency path including the resistor R1dominates, however the resistor R1 does attenuate the low frequencysignals. As shown in FIG. 7c , the phase response is relatively flat.

FIG. 8 shows a gain compensated high pass lead-lag network 80. The gaincompensated high pass lag-lead network 80 comprises a low frequency path82 and a high frequency path 84. The low frequency path 82 and the highfrequency path are connected in parallel between an input terminal 83and an output terminal 85. The high frequency path 84 comprises acapacitor C5 which is the same as the high frequency path 74 of thelead-lag network 71 shown in FIG. 7b . The low frequency path 82comprises an amplifier G2 connected in series with a first resistor R1.A second resistor is connected between the output terminal 85 andground.

The amplifier G2 compensates for the decreased gain of the low frequencypath and allows the low frequency gain to be matched to the highfrequency gain. The gain of G2 depends on the ratio of R1 to R2 whichare set according to the following formula if equal gain is required forboth high and low frequencies:

$\frac{G\; 2 \times R\; 2}{{R\; 1} + {R\; 2}} = 1$

The gain compensated lag-lead network 80 has a flat gain and phaseresponse.

FIG. 9a shows an envelope modulator 90 according to an embodiment. Theenvelope modulator 90 is based on the gain compensated high passlead-lag network 80 shown in FIG. 8. The envelope modulator 90 comprisesa low frequency path 92 and a high frequency path 94 which are connectedin parallel between an input terminal 93 and an output load R2.

The low frequency path 92 comprises a first resistor R4, a switched modepower supply SMPS, a second resistor R1 and an inductor L5 connected inseries. A capacitor C4 is connected between the input of the switchedmode power supply SMPS and ground. The capacitor C4 and the firstresistor R4 form a low pass filter.

The switched mode power supply SMPS includes a gain G2 to compensate forthe presence of the second resistor R1 in the low frequency path. Thusin the envelope modulator 90 shown in FIG. 9a the switched mode powersupply SMPS incorporates the gain G2 shown in the gain compensated highpass lag lead network 80 shown in FIG. 8.

The high frequency path 94 comprises an amplifier G1 and a capacitor C5connected in series.

Adding the amplifier G1 to the high frequency path and the inductor L5and the low pass filter formed by R4 and C4 to the low frequency pathhas little effect on the frequency response. As discussed above inrelation to FIG. 4a , the values of the components are selected so thatthe cut off frequency of the high pass filter formed by C5 and R2 islower than the cut off frequency of the filter formed by R1, L5 and R2.For example, the cut off frequency of C5 and R2 could be 5 times lowerthan that of R1, L5 and R2.

FIG. 9b shows the frequency response of the envelope modulator 90 shownin FIG. 9a . As discussed above in relation to FIG. 4b , in FIG. 9b itis assumed that R1=R2. In this case as shown in FIG. 9b there is a dipin gain of −7 dB at the cross-over frequency and there is a phasedistortion from −25 degrees to +25 degrees around the cross-overfrequency.

Here it is assumed for simplicity that R1=R2. If R1 is equal to R2, thegain distortion is 7 dB and the phase distortion is 50°. However thisdoes not have to be the case and different values for R1 and R2 may beused.

As discussed above in relation to FIG. 5, the efficiency can beincreased by replacing the resistor R1 with a synthetic impedance.

FIG. 10 shows an envelope modulator 100 according to an embodiment. Theenvelope modulator 100 is based on the envelope modulator 90 shown inFIG. 9a but the resistor R1 is replaced by a synthetic impedance formedby a resistor R6, an amplifier G2 and the positive and negativefeedback.

The envelope modulator 100 comprises a low frequency path 102 and a highfrequency path 104 connected in parallel between an input terminal 103and an output load R2. The high frequency path 104 is the same as thehigh frequency path 94 of the envelope modulator 90 shown in FIG. 9 a.

The low frequency path 102 comprises a first resistor R4, a switchedmode power supply SMPS, the resistor R6 and an inductor L5 connected inseries. The switched mode power supply SMPS includes a gain G2 tocompensate for the presence of the resistor R6. A capacitor C4 isconnected between the input of the switched mode power supply SMPS andground. The capacitor C4 and the first resistor R4 form a low passfilter.

Two feedback loops are connected to the input of the switched mode powersupply SMPS. A first feedback loop 107 is connected to the output of theamplifier before the resistor R6. The feedback signal of the firstfeedback loop is subtracted from the input of the switched mode powersupply SMPS. A second feedback loop 108 is connected to a point betweenthe resistor R6 and the inductor L5. The feedback signal of the secondfeedback loop is added to the input to the switched mode power supplySMPS. The synthetic impedance may be realised using the circuit shown inFIG. 6.

The two cut-off frequencies of the envelope modulators of embodimentsare designed to overlap. The performance reported here is for whenR1=R5. This will halve the efficiency of whichever path (AC or DC) it isplaced in and reduce overall system efficiency. Reducing the value of R1relative to R5 increases the magnitude of the phase transient at thecrossover frequency. High impedance can be synthesised from R6 as shownin FIG. 5 and FIG. 10 to maintain the linearity, without the loss inefficiency.

FIG. 11 shows a power amplifier circuit 1 according to an embodiment.The power amplifier circuit comprises an input terminal 2, an envelopedetector 3, an envelope modulator 4, a power amplifier 5 and an outputterminal 6. The envelope modulator 4 comprises a lag-lead network or alead-lag network as described above.

In use, a radiofrequency signal to be amplified is applied to the inputterminal 2. The envelope detector 3 detects an envelope signal from theradiofrequency input signal. The envelope modulator 4 amplifies theenvelope signal and modulates the power supply to the power amplifier 5according to the envelope signal. The power amplifier 5 amplifies theradiofrequency signal and operates in saturation mode with the powersupply voltages setting the envelope of the amplified radiofrequencysignal supplied to the output terminal 6.

Modern communication and broadcast systems require a low transmittererror vector magnitude. It has been found that the phase transients suchas those shown in FIG. 1b have a higher impact on error vector magnitudethan any distortion in the gain response. Hence it is more important tocorrect for the phase distortion than for any gain distortion.Embodiments described herein allow for the correction of the phasedistortion and therefore a large reduction in error vector magnitudewith a small number of extra components.

Embodiments described herein have applications in transmitters such asLTE base stations and DVB-T2 transmitters.

When simulated with a 3 MHz bandwidth LTE signal, it has been found thatthe envelope modulators described herein can almost complete removephase flip and have a resulting error vector magnitude of 0.91%. Theenvelope modulators described herein have also been found to improveadjacent channel power ratio (ACPR) by approximately 10 dB.

While certain embodiments have been described, these embodiments havebeen presented by way of example only, and are not intended to limit thescope of the inventions. Indeed the novel circuits described herein maybe embodied in a variety of other forms; furthermore, various omissions,substitutions and changes in the form of methods and apparatus describedherein may be made without departing from the spirit of the inventions.The accompanying claims and their equivalents are intended to cover suchforms of modifications as would fall within the scope and spirit of theinventions.

1. An envelope modulator comprising a gain compensated lag-lead networkor a gain compensated lead-lag network.
 2. An envelope modulatoraccording to claim 1 comprising a low frequency path configured toamplify components of an input signal having frequencies below a firstthreshold and a high frequency path configured to amplify components ofthe input signal having frequencies above a second threshold.
 3. Anenvelope modulator according to claim 2 wherein the first threshold isgreater than the second threshold.
 4. An envelope modulator according toclaim 3 wherein the first threshold is at least five times greater thanthe second threshold.
 5. An envelope modulator according to claim 2,wherein the modulator comprises a lag-lead network and the highfrequency path comprises a first impedance and a first amplifierconfigured to provide gain compensation for the first impedance.
 6. Anenvelope modulator according to claim 5, wherein the first impedance isa synthetic impedance.
 7. An envelope modulator according to claim 2,wherein the modulator comprises a lead-lag network and the low frequencypath comprises a second impedance and a second amplifier configured toprovide gain compensation for the second impedance.
 8. An envelopemodulator according to claim 7, wherein the second impedance is asynthetic impedance.
 9. An envelope modulator according to claim 7,wherein the second amplifier is a switched mode power supply.
 10. Apower amplifier circuit comprising a power amplifier configured toamplify an input radiofrequency signal and an envelope modulatoraccording to claim 1 configured to modulate a power supply of the poweramplifier according to an envelope of the input radiofrequency signal.